Modified full bridge dual inductive coupling resonant converter for electric vehicle battery charging applications

Received Jul 23, 2021 Revised Feb 28, 2022 Accepted Mar 10, 2022 In this paper, a modified full bridge dual inductive coupling (LCL) resonant converter for electric vehicles (EVs) battery charging applications is proposed. The main objective of the proposed topology is to operate the converter in constant voltage (CV) and constant current (CC) mode during battery charging. The presented topology's uniqueness comprises the following: i) isolated charging and power factor correction (PFC), ii) to achieve zero-voltage switching (ZVS) and zero-current switching (ZCS) for inverter switches, iii) reduction of number of rectifier diodes to reduce the conduction and switching losses, and iv) reducing the magnetizing current. The output voltage dependence of resonant converter is reduced using a PFC converter against the variations of the alternating current (AC) grid input voltage. The variations of the wide range output voltage and load is compensated by a small variation in switching frequency. The proposed topology's detailed operation is simulated using the MATLAB/Simulink tool.


INTRODUCTION
The demand for electric vehicles (EVs) and plug-in hybrid electric vehicles (PHEVs) is ever increasing due to an impending scarcity of fossil fuels is expected. and the zero emission of carbon dioxide (CO2), noise reduction, smooth functioning, and high efficiency. The power electronic converters in EVs and PHEVs play a significant role in interfacing the battery and grid to drive the traction motor which are discussed in [1]- [5]. The design of reliable, compact and efficient power electronic converters is of great interest and challenge for the researchers and engineers [6]- [8].
Presently, many topologies have been proposed in the literature for isolated battery charging using inductors capacitor (LLC) resonant converters [9]- [22]. The topologies proposed in reference [9]- [17] describes the operation of direct current/alternating current (DC/AC) and AC/DC converter for battery charging with high frequency transformer to achieve isolation during charging. Hu et al. [9] a modified LLC converter with two transformers in series is proposed for wide input voltage range. Based on the input voltage, a scheme is developed to reduce the magnetizing current by modifying the magnetising inductance adaptively while keeping high DC voltage gain. However, the topology needs two extra switches and four diodes which eventually leads to conduction loss as well as control complexity in operating extra switches. Dusmez and Khaligh [10], by incorporating the resonant inductor inside the transformer, the LLC resonant converter's bulk is reduced, making it particularly appropriate for use as an integrated onboard charger for electric vehicles. However, the report gives no indication of the imbalanced leakage inductance that is fairly prevalent in centre tapped transformers with bifilar windings or the parameter changes that occur when a resonant inductor is used. Wu et al. [11] proposes a unique secondary-side phase-shift-controlled LLC resonant converter to decrease magnetising inductance circulating current. The topology is not designed for high DC voltage gain, thus activities like constant voltage (CV) charge, constant current (CC) charge, zerovoltage switching (ZVS), and zero-current switching (ZCS) are not reported. To eliminate low and high frequency current ripple on the battery a new resonant tank for an LLC resonant DC-DC converter is proposed in [12] considering only half bridge inverter which doesn't guarantee high DC voltage gain. Though, peak to peak low frequency voltage ripple of 0.14 V is achieved but the peak to peak low frequency current ripple is limited to 0.537 A only. An LLC resonant DC-DC converter design procedure is presented in reference [13] for an onboard lithium-ion battery charger of a PHEV. Based on fundamental harmonic approximation (FHA) the worst-case circumstances for primary-side ZVS operation are identified analytically and charging profile is implemented considering a constant maximum power (CMP). To ensure soft-switching throughout the operation, worst-case operating point is used. To avoid the inaccuracy of FHA approach below the resonance region, based on specific operation mode the design constraints are explained briefly. In order to validate the designed equations only full bridge LLC resonant converter is considered which doesn't guarantee high DC voltage gain or high DC current gain. Deng et al. [14], a dual-bridge LLC resonant converter for wide input voltage range is proposed and a fixed frequency pulse width modulation (PWM) control scheme is employed. The magnetising inductor limits the voltage gain range regardless of the quality factor. In addition to this, two extra switches in the topology increases the cost and control complexities. An LLC resonant converter is proposed in reference [15] with considerable resonant inductance in order to form an adjustable wide-range regulated voltage source. Soft switching is employed by properly choosing the dead time and maximum switching frequency for power switches under inductive region operating conditions. A frequency control strategy is used for the proposed topology to handle wide regulated output voltages against wide input voltage or output load variations. A novel dual full-bridge LLC (FBLLC) resonant converter for implementing CC and CV charges is proposed in reference [16]. The paper quickly explains ZVS for all primary switches in CV charge and approximately ZCS for all primary switches in CC charge. Both CC and CV charge consider fixed frequency resonant operation to limit circulating current. To get high DC voltage gain, the article includes an extra diode bridge, and which results in greater conduction losses. However, the proposed topology is claimed to be operated at fixed switching frequency for both CC and CV operations, although clarity on how same switching frequency may be used for two different resonant frequencies is lacking. This is significant because when the resonance frequency is lower than the switching frequency, the active switches of the inverter suffer from excessive turn-off current, when it is greater, and the main side of the transformer suffers from circulating current. To operate a lower resonant converter in CC charge mode, turn off the switch connected in series with an extra capacitor, and turn on the switch to lower the resonant frequency. This contradicts the paper's CC and CV charge operational point of view. Moreover, while the switch is off, the circuit lacks a discharge pathway for the added capacitor, resulting heating and a deficit of current restriction. Besides the aforesaid, the suggested topology's closed loop control approach lacks clarity on selecting a common proportional-integral (PI) controller for both CV and CC charges. The paper also fails to describe the resonant converter's output voltage dependence on the ac grid input voltage.
In the present paper, a modified full bridge LCL resonant converter topology is proposed to address the afore-mentioned issues. The proposed topology comprises of power factor correction (PFC) converter and an isolated full bridge dual-inductive coupling converter (FBD-LCL) is briefly explained in the next section. The organization of the paper is being as: in section 2, the description of the proposed topology is explained along with various modes of operation. Section 3 discusses, the control techniques used for the proposed topology. In section 4 design analysis are included. The simulation results are included in section 5 and the conclusion are outlined in section 6.

DESCRIPTION OF THE PROPOSED TOPOLOGY
The proposed modified FBD-LCL converter topology is shown in Figure 1 includes an active switch Sb, an inductor L1, a diode D5 and a capacitor C2 in PFC converter, four active switches (S1-S4) constitutes full bridge inverter with body diodes (

Power factor correction converter
The PFC converter is the first stage in EVs and PHEVs charging system designed to provide power factor correction. As per international electrotechnical commission (IEC) 1000-3-2 standard, it is mandatory to limit the harmonic current drawn by the charging system to low, which is measured through total harmonic distortion (THD) and to generate input current waveform close to sinusoidal. In addition to this, when the converter operates at high switching frequency to achieve higher power density, increasing the efficiency, and reducing the electro-magnetic interference (EMI) is a major concern. The EMI standard EN55022 class B formed to control and eliminate high frequency radiated and conducted EMI which can inject unwanted signals into the neighboring circuits and/or systems [17]- [19]. In order to fulfill the requirements of the above standards, boost converter is used as PFC converter in the proposed topology due to its simplicity and reduced size of the filter. The PFC converter, in addition to perform power factor correction function it also provides constant DC voltage to the full bridge inverter.

Operating principle of the proposed converter
The proposed converter consists of two LCL resonant converters, upper resonant converter FBD-LCL1, and lower resonant converter FBD-LCL2. The operating principle of proposed converter is explained through full bridge LCL resonant converter based on FHA method as shown in Figure 2 to Figure 5. Each LCL resonant converter operates at two different resonant frequencies, upper resonant frequency ( ) and lower resonant frequency ( ). The resonant converter is operated as constant voltage source at as in (1) and is operated as constant current source at as in (2). (1) When the resonant converter operates at it behaves as a constant voltage source, the DC voltage gain is obtained based on FHA analysis [6] is given by (3): Where = DC voltage gain. Normalized frequency = , is switching frequency. Inductance ratio = is the output DC resistance. is the transformer turns ratio. When the resonant converter operates at it behaves as a constant current source, the output current is obtained as in (4): where = 2 , = .
The graph between voltage gain (GV) versus normalized frequency (fx) with constant inductor ratio (Li) of 5 and variable quality factor (Q) is shown in Figure 2. It is very clear from the graph that, as Q increases GV decreases, at fxmin gain is highest at fxmax gain is lowest and at fx = fs gain is unity. The ZVS and ZCS regions are also indicated in the graph. In Figure 3, Gain versus normalized frequency with constant quality factor and current increases at fx = fs for increasing value of Q and decreases when fx<1 or fx>1.

Charge and discharge profile characteristics of the battery
The charge and discharge profile of voltage, current, power and equivalent impedance of EV battery in CC and CV mode are given in Figure 6 and Figure 7 respectively. The voltage varies from a minimum voltage of 220 V to a maximum voltage of 400 V for Li-ion battery which consists of 96 cells. The battery charging operation starts with CC mode with a constant current of 2.5 A and ends when the battery attains a maximum voltage equal to 400 V. Using mode selector switch Sr, the battery charging operation is changed from CC mode to CV charge mode. In CV charge mode, the battery voltage is maintained at constant voltage of 400 V but the current drawn by the battery is decreased from an initial current of 2.5 A to final current of 0.25 A at the end of this mode. It is also clear from charge profile characteristics indicated in Figure 6 (leftfigure) that, the FBD-LCL1 converter is designed to provide constant voltage of 200 V, whereas FBD-LCL2 converter is designed to provide an initial voltage of 20 V in the beginning of CC charge mode and a final voltage of 200 V at the end of CC charge mode. Both FBD-LCL1 and FBD-LCL2 converters are designed to deliver a continuous 200 V to get a battery voltage of 400 V. Figure 6 (right) shows the battery power and impedance during CC/CV charge mode. The initial battery power is 550 W and the final battery power is 1000 W. At the end of CV charge mode, it drops to 100 W. The starting battery impedance is 88 and reaches 160 at the conclusion of CC charge mode, then gradually climbs as the battery current demand drops. The discharge profile characteristics of EV battery is also included in Figure 7, left-figure indicates battery voltage and current and right-figure indicates battery power and impedance during battery discharge.

Constant current charge mode
In this mode upper resonant converter FBD-LCL1 behaves as constant voltage source and lower resonant converter FBD-LCL2 behaves as constant current source. The upper resonant converter FBD-LCL1 resonate at two frequencies, upper resonant frequency 1_ as in (5) and lower resonant frequency 1_ as in (6).
In order to operate FBD-LCL1 converter as constant voltage source, it is operated at upper resonant frequency 1_ . The lower resonant converter FBD-LCL2 resonate at two frequencies, upper resonant frequency 2_ as in (7) and lower resonant frequency 2_ as in (8).  as in (11) and lower resonant frequency 2_ as in (12).
In order to operate FBD-LCL2 converter as constant voltage source, it is operated at upper resonant frequency 2_ .

CONTROL TECHNIQUE
The control technique for the proposed topology to obtain DC link voltage and to charge battery are shown in Figure 8 and Figure 9 respectively. The references [23]- [25] deliberates few governing control techniques used in EV applications for power factor correction. The implementation of CC and CV charge mode of operation needs closed loop control scheme. Hence, the external feedback signals from the DC link voltage, battery voltage, battery current, PFC inductor current, and resonating inductor current are used to choose the control method inputs for the proposed design. The PFC converter and FBD-LCL converter control techniques are concisely presented in the following sections.

Power factor correction converter control technique
In Figure 9, the PFC converter is controlled by a PI controller, rectifier, adder/subtractor, and comparator. A transformer and a resistive network are used to reduce the AC grid voltage to the desired level. This AC voltage is rectified and filtered to produce Vr, the reference DC voltage. The error signal is obtained by comparing the reference Vbref to the observed Vb. For the regulated step signal, this error signal is sent into the PI controller's reference current input Ir. The step signal is compared to the high frequency saw tooth signal to generate a PWM signal for the switch Sb.

Full bridge dual-inductive coupling converter converter control technique
The Figure 9 shows three proportional integral derivative (PID) controllers, a mode selector switch, three subtractors, a limiter, a voltage-controlled oscillator (VCO), and a comparator. As long as the battery voltage is below the peak charge voltage, the mode selector switch will activate the current controller to charge the battery. When the battery voltage hits 400 V, the mode selector switch automatically shifts from CC to CV. The error signal from the inner loop PID controller is supplied to the mode selector switch output. The step signal is limited for saturation and then sent to a VCO for variable frequency. The PWM pulses for MOSFET switches S1, S2, S3, and S4 have configurable switching frequency. Simulink auto tuner finds Kp, Ki, and Kd values for all three PIDs.

DESIGNING PARAMETERS AND EQUATION
The factors needed to select switches, diodes, resonant circuits whose primary goal is to achieve soft switching conditions across its operating range and passive components are discussed in this section.

Selection of switches and diodes
The collection of a power semiconductor switches and diodes is based on reverse blocking voltage, maximum forward current, and power handling capabilities. In the proposed topology the inverter module consists of five MOSFET switches Sb, S1, S2, S3, and S4 with body diodes and the bridge rectifier circuit is formed with nine additional diodes D1-D5, and D10-D13 are considered.

Selection of transformer turns ratio
The transformer turns ratio is defined as: where is rated input voltage, is rated output voltage and is the forward voltage drop of rectifier diode.

Selection of resonating elements and magnetizing inductances
The selection of resonating elements inductance, capacitance and magnetizing inductances which are used in Figure 1 for the upper resonant converter elements and the lower resonant converter elements are calculated based on constant current charge and constant voltage charge mode. The equations pertaining to the calculations of these inductances and capacitances are included in (5) to (12).

Selection of rectifier, direct current-link and battery side capacitors
Three capacitors C1 at the output of the input bridge rectifier, C2 at the high DC-link, and C3 across the battery are used in the proposed converter. The formulas for constructing these capacitors are as shown in:

RESULTS AND DISCUSSION
The results and discussion of the proposed topology are briefly explained in this section. The simulation of the proposed topology was verified using MATLAB/Simulink version R2019a for the Table 1 specifications. Figures 10 to 14 show the simulation results for power factor correction, switching pulses, inverter voltage and current, ZVS and ZCS operation, transformer primary and secondary voltage and current, and voltage and current through rectifying diodes. To produce the rated output voltage of 400 V, the rectifier stage generates a 380-420 V DC voltage at the output terminal.
The switching and resonant frequencies are 50 kHz. VCr is the resonant capacitor voltage, VDS1 is the drain to source voltage of S1, IDS1 is the drain to source current via S1, vD10, and vD12 are voltages and iD10 and iD12 are currents of rectifier diodes D10 and D12 respectively. The transformer primary voltage, primary current, secondary voltage and secondary current are vP1, iP1, vS1, iS1 respectively. Figure 10 shows the grid voltage, grid current, DC link voltage, and DC link current waveforms. The measured power factor (PF) is 0.99 with a lower current harmonic distortion (ITHD) of 0.97. Figure 11 shows that ZVS of power switches and ZCS of rectifier diodes were achieved at 400 V output voltage. In Figure 12 switching pulse between gate and source of S1, drain to source voltage of S1 and current through S1 switch indicating ZVS turn-on and ZCS turn-off conditions are shown. It is very clear from the results that when voltage is zero switch is turned-off and when current is zero switch is turned-on. The transformer primary voltage, primary current, secondary voltage and secondary current are given in Figure 13 and in Figure 14, voltage across and current through rectifier diode D10 indicating ZVS turn-on and ZCS turn-off are included. Similar operations can also be performed on other rectifying diodes (D11, D12, and D13) in the secondary side full-bridge rectifier.  Lr and Lm and current through rectifier diodes iD10 and iD12 Figure 12. Waveforms of switching pulse applied between gate and source of S1, drain to source voltage of S1 and current through S1 switch indicating ZVS turn-on and ZCS turn-off

CONCLUSION
In this paper a modified full bridge dual LCL resonant converter fit for implementing both CC and CV charge operation for charging the batteries is proposed. The PFC converter is also included in the proposed topology for power factor correction. The implementation of two resonant converters to resonate at lower and upper resonant frequencies, results in ZVS and ZCS of the converter switches in the CC charge and CV charge operations. The ZVS turn-on and ZCS turn-off is also achieved during rectification operation of the converter at the output. The higher resonant frequency is obtained by operating lower resonant converter also in CV charge mode with the help of mode selector switch. The operation of ZVS and ZCS ensure that the converter operates with no switching losses. This will improve the efficiency of the converter over a wide range of output voltage and hence the suggested architecture catches instant applications in battery charging systems of electric vehicles.